Path search circuit dividing a received signal into a plurality of FFT windows to reduce arithmetic operation processes for cross-correlation coefficients

ABSTRACT

A path search circuit wherein a received signal is divided with a plurality of FFT windows to reduce arithmetic operation processing for cross-correlation coefficients is disclosed. An interleave unit divides received signal rxd into two rxd 1 , rxd 2  at one-chip intervals, and the two sequences are picked out with overlapped FFT windows and FFT is performed for the picked out sequences by two FFT units. A cross power spectrum calculating unit determines cross power spectra between the received signal after the FFT and a reference signal stored in a reference signal storage unit. An output of the cross power spectrum calculating unit is averaged for each FFT window by an averaging unit, and IFFT is performed for the averaged cross power spectra by an IFFT unit. The two resulting cross-correlation coefficients are rearranged in order of time by a deinterleave unit and interpolated to an accuracy necessary for detection of a path timing by an interpolation unit.

BACKGROUND OF THE INVENTION

[0001] 1. Field of the Invention

[0002] This invention relates to a mobile telephone or portabletelephone system (cellular system) which uses a direct sequence codedivision multiple access (DS-CDMA) communication method, and moreparticularly to a path search circuit for a base station radioapparatus.

[0003] 2. Description of the Relates Art

[0004] Mobile communications systems in which a CDMA communicationmethod is used have been developed in recent years and include systemswhich are based on the IS-95 standards (TIA/EIA) and have been put intopractical use already and W-CDMA (Wideband Code Division MultipleAccess) systems which are third generation mobile communications systemswhose standardization is being proceeded in the 3 GPP (3rd GenerationPartnership Project) although they have not been put into practical useas yet.

[0005] In a system based on the IS-95 standards, a spread code obtainedby multiplying a PN code having a comparatively long period of 26.6 ms(80 ms/3, 32768 chips) and a Walsh code of a code length of 64 is usedas a spread code for a downlink which is a link from a base station to amobile station. As the PN code, different codes (accurately, codesshifted by predetermined number of times from the same spread code) areused by different base stations and, even in the same base station, fordifferent sector antennae. The Walsh code of the code length of 64 isused to distinguish a plurality of channels transmitted from one sectorantenna (for the CDMA, the same carrier is shared by a plurality ofchannels and the channels are distinguished with the spread code). Apilot channel which is not modulated with data is transmitted with acomparatively high power for each sector. For the Walsh code used in thepilot channel, the 0th code, i.e., a code of all “0”s, is used. In otherwords, a signal transmitted by the pilot channel is a predetermined codesequence having a period of 26.6 ms. Accordingly, a mobile station of asystem which is based on the IS-95 standards uses the pilot channel todetect a peak of a cross-correlation between a predetermined codesequence of the pilot channel and a received signal to detect a pathtiming. The period of the spread code is 32,768 chips and is too long todetermine cross-correlation coefficients at a time. Therefore, a slidingcorrelator is used to successively determine correlation coefficientswhile the received signal and a reference signal (the predeterminedspread code of the pilot channel) are successively shifted in time.

[0006] A conventional reception timing detection method (chipsynchronization) is disclosed in the following reference document 1, forexample:

[0007] Reference document 1: Andrew J. Viterbi, “Principle of SpreadSpectrum Communication”, April 1995, Chapter 3, pp. 39-66, FIGS. 3.1,3.2 and 3.6

[0008] According to the reception timing detection method disclosed inthe reference document 1, acquisition of a timing of a signal spreadwith a spread code, which is a pseudo random code, is performed in twostages. In particular, the acquisition is divided into two stages ofinitial synchronization acquisition (search) and synchronizationtracking (tracking). An initial synchronization acquisition (search)method is a method of serially searching for a timing while thereception timing is successively displaced by a ½ chip interval untilthe correlation power exceeds a certain threshold value as recited inChapter 3, Paragraph 4 of the reference document 1. The synchronizationtracking (tracking) is performed by a method called early-late gate orDLL (Delay Lock Loop). According to the methods, a correlation power ata timing earlier by Δt of a delay time than a timing at which the signalis to be received and another correlation power at another timing laterby Δt are determined, and the timing is finely adjusted so that thedifference between the correlation powers may be reduced to 0.

[0009] After a path timing is detected once, it is only required that avariation of the path timing can synchronously follow up (tracking) thevariation of the propagation time between a base station and a mobilestation which is caused by a movement of the mobile station and thevariation of the propagation time caused by a positional relationshipwith a reflecting object or objects in multi-path propagation paths.Therefore, a cross-correlation coefficient (which represents a delay4profile of the propagation path) may be determined within the range ofseveral microseconds to several tens microseconds before and after thepresent timing. Determination of a cross-correlation coefficient (delayprofile) within the range limited in this manner can be realized also bya plurality of correlators which operate simultaneously. Where it can beregarded that the path timing varies continuously in time, also a timingfollow-up method in which a DLL is used as disclosed in theaforementioned reference document 1 has been realized.

[0010] Various standards for the W-CDMA are recited in the followingreference document 2:

[0011] Reference document 2: 3rd Generation Partnership Project;Technical Specification Group Radio Access Network; Spreading andmodulation (FDD), 3G TS 25.213 version 3.1.0, December 1999

[0012] According to the reference document 2, the W-CDMA uses a Goldcode of a period of 10 ms as a spread code and a Walsh code of a periodof 1 symbol (the code length varies depending upon the symbol rate). Inthe downlink, different Gold codes are used in different base stationsand for different sectors in the same base station. In the uplink from amobile station to a base station, Gold codes different among differentmobile stations are used, and different Walsh codes are allocated todifferent physical channels in the same mobile stations. In both of theuplink and the downlink, a pilot symbol modulated with a predeterminedcode sequence is multiplexed (code multiplexed and time multiplexed).

[0013] Different from the downlink in a system based on the IS-95standards, a pilot symbol of the W-CDMA is not spread with the samespread code (including shifted spread codes) among all base stations orall mobile stations. However, if the spread code is known, then a codesequence of the pilot symbol can be considered as a completely knowncode sequence. Accordingly, in the W-CDMA, a path timing can be detectedby using the pilot symbol to detect a peak of a cross-correlationbetween a predetermined code sequence of the pilot symbol and a receivedsignal. As a conventional path timing detection method for the W-CDMA, a“Reception Timing Detection Circuit for a CDMA Receiver” disclosed inJapanese Patent Laid-Open No. 32523/1998, for example, is known.

[0014] In the W-CDMA, however, since the chip rate is higher than thatof a system based on the IS-95 standards, a variation of the multi-pathpaths frequently varies the path timing discontinuously for more thanone chip interval (because one chip is 60 ns, it corresponds to apropagation path difference of approximately 78 m). Accordingly,sufficient path tracking cannot be achieved by means of a DLL or a likescheme that is conventionally used for synchronization tracking(tracking) in a system based on the IS-95 standards or a like system.

[0015] Meanwhile, a path search circuit which uses the conventional pathtiming detection method has a problem in that, where the spread ratio ofa pilot symbol is high and the code length of the pilot symbol is great,if a cross-correlation coefficient is calculated in the time domain,then a very great amount of calculation is required. Furthermore, a basestation must detects path timings of received signals from a pluralityof mobile stations, and the prior art has another problem in that anumber of identical circuits equal to the number of mobile stations mustbe prepared and a great amount of calculation is required.

SUMMARY OF THE INVENTION

[0016] It is an object of the present invention to provide a path searchcircuit which can reduce the amount of arithmetic operation required forpath search and path tracking in a mobile communications system(cellular system) that uses a DS-CDMA communication method.

[0017] It is another object of the present invention to provide a pathsearch circuit which, when applied to a base station apparatus of acellular system which uses a DS-CDMA communication method, can reducethe amount of arithmetic operation required for path search and pathtracking of received signals from a plurality of mobile stations.

[0018] According to the present invention, a path search circuit for areceiver which uses a DS-CDMA communication method is for detecting apath timing, which is a timing at which spreading is performed on thetransmission side, from a received radio signal and comprises a radioreceiving unit, an A/D converter, a cross-correlation coefficientcalculating unit, a cross-correlation coefficient averaging unit, and apeak detecting unit.

[0019] The radio receiving unit filters and frequency converts thereceived radio signal to convert the received radio signal into abaseband signal. The A/D converter samples the baseband signal at asampling rate equal to N times a chip rate to convert the basebandsignal into a digital signal.

[0020] The cross-correlation coefficient calculating unit includesinterleave means, N fast Fourier transform means, reference signalstorage means, N cross power spectrum calculating means, N inverse fastFourier transform means, and deinterleave means.

[0021] The interleave means rearranges the baseband signal digitized bythe A/D converter into N sequences sampled at chip intervals. The N fastFourier transform means pick out the N received signal sequencesrearranged by the interleave means with mutually overlapped FFT windowsof a predetermined time length and performs fast Fourier transform forthe picked out received signal sequences. The reference signal storagemeans stores a signal sequence produced by picking out a predeterminedcode sequence with FFT windows of a fixed time length and fast Fouriertransforming the picked out code sequence as a reference signal.

[0022] The N cross power spectrum calculation means determine theproduct of the received signal fast Fourier transformed by the fastFourier transform means and a complex conjugate number of the referencesignal stored in the reference signal storage means for each of the FFTwindows to determine cross power spectra between the received signal andthe predetermined code sequence.

[0023] The N cross power spectrum averaging means averages the crosspower spectra for the respective FFT windows. The N inverse fast Fouriertransform means inverse fast Fourier transform the N cross power spectraaveraged by the cross power spectrum averaging means to convert thecross power spectra into N cross-correlation coefficients and output theN cross-correlation coefficients.

[0024] The deinterleave means rearranges the N cross-correlationcoefficients output from the respective inverse fast Fourier transformmeans in order of time to produce and output a single cross-correlationcoefficient.

[0025] The cross-correlation coefficient averaging unit averages thecross-correlation coefficient output from the cross-correlationcalculating unit over a fixed period of time. The peak detecting unitdetects one or a plurality of peaks from the cross-correlationcoefficient averaged by the cross-correlation coefficient averaging unitand outputs a timing at which the peak or each of the peaks is obtainedas a path timing.

[0026] According to an embodiment of the present invention, in the pathsearch circuit for a receiver which uses a DS-CDMA communication method,the value N may be an integer lowest among values which satisfy therelationship: N≧radio bandwidth/chip rate.

[0027] According to another embodiment of the present invention, thepath search circuit may further comprise interpolation means foroversampling the N cross-correlation functions output from thedeinterleave means as sequences of a time interval equal to 1/N the chipinterval to M times, where M is a positive integer, and passing theoversampled cross-correlation coefficients through a low-pass filter toproduce cross-correlation coefficients oversampled to N×M times the chiprate and then outputting the produced cross-correlation coefficients.

[0028] According to another aspect of the present invention, a pathsearch circuit for a receiver which uses a DS-CDMA communication methodcomprises a radio receiving unit, an A/D converter, a cross-correlationcoefficient calculating unit, a cross-correlation coefficient averagingunit, and a peak detecting unit.

[0029] The radio receiving unit filters and frequency converts thereceived radio signal to convert the received radio signal into abaseband signal. The A/D converter samples the baseband signal at asampling rate equal to N times a chip rate to convert the basebandsignal into a digital signal.

[0030] The cross-correlation coefficient calculating unit includesinterleave means, N fast Fourier transform means, reference signalstorage means, N cross power spectrum calculation means, N cross powerspectrum averaging means, N first cross power spectrum conversion means,cross power spectrum addition means, second cross power spectrumconversion means, and inverse fast Fourier transform means.

[0031] The interleave means rearranges the baseband signal digitized bythe A/D converter into N sequences sampled at chip intervals. The N fastFourier transform means pick out the N received signal sequencesrearranged by the interleave means with mutually overlapped FFT windowsof a predetermined time length and perform fast Fourier transform forthe picked out received signal sequences. The reference signal storagemeans stores a signal sequence produced by picking out a predeterminedcode sequence with FFT windows of a fixed time length and fast Fouriertransforming the picked out code sequence as a reference signal.

[0032] The N cross power spectrum calculation means determine theproduct of the received signal fast Fourier transformed by the fastFourier transform means and a complex conjugate number of the referencesignal stored in the reference signal storage means for each of the FFTwindows to determine cross power spectra between the received signal andthe predetermined code sequence. The N cross power spectrum averagingmeans average the cross power spectra for the respective FFT windows.The N first cross power spectrum conversion means apply reflection by Ntimes and phase rotation in the frequency domain to the N cross powerspectra averaged by the cross power spectrum averaging means and havinga bandwidth equal to the chip rate to convert the N cross power spectrainto a single cross power spectrum having a bandwidth equal to N timesthe chip rate.

[0033] The cross power spectrum addition means adds the N cross powerspectra converted by each of the first cross power spectrum conversionmeans. The second cross power spectrum conversion means adds the numberof “0”s equal to N×(M−1) times the chip rate to a high frequency of thecross power spectrum obtained by the addition means, where M is apositive integer. The inverse fast Fourier transform means inverse fastFourier transforms the cross power spectrum obtained by the second powerspectrum conversion means and having a bandwidth increased to M times todetermine a cross-correlation coefficient.

[0034] The cross-correlation coefficient averaging unit averages thecross-correlation coefficients output from the cross-correlationcalculating unit over a fixed period of time. The peak detecting unitdetects one or a plurality of peaks from the cross-correlationcoefficient averaged by the cross-correlation coefficient averaging unitand outputs a timing at which the peak or each of the peaks is obtainedas a path timing.

[0035] According to the present invention, an advantage can beanticipated that arithmetic operation processing of cross-correlationcoefficients essentially required for path search can be reduced bydividing a received signal into a plurality of FFT windows to performfast Fourier transform and performing multiplication by a referencesignal and averaging in the frequency domain. Particularly since thenumber of chips per FFT window can be increased by dividing a receivedsignal into a plurality of sequences at one-chip intervals to performfast Fourier transform, the overlap between FFT windows can be reduced.Since this decreases the number of FFT windows per received signal for afixed time length, an advantage can be anticipated that the amount ofarithmetic operation can be reduced.

[0036] If it is desired to raise the accuracy of a path timing higherthan a sampling period of the A/D converter, A/D conversion is performedat the possible lowest sampling rate and cross-correlation coefficientsor cross power spectra are determined, and then interpolation isperformed in the time domain with a required time accuracy. This makesit possible to suppress the amount of arithmetic operation in fastFourier transform operation, which involves a very great amount ofarithmetic operation, to the possible lowest amount.

[0037] According to a further aspect of the present invention, a pathsearch circuit for a receiver which uses a DS-CDMA communication methodcomprises a radio receiving unit, an A/D converter, a cross-correlationcoefficient calculating unit, a cross-correlation coefficient averagingunit, and a peak detecting unit.

[0038] The radio receiving unit filters and frequency converts thereceived radio signal to convert the received radio signal into abaseband signal. The A/D converter samples the baseband signal at asampling rate equal to N times a chip rate to convert the basebandsignal into a digital signal.

[0039] The cross-correlation coefficient calculating unit includesinterleave means, N fast Fourier transform means, reference signalstorage means, cross power spectrum calculation means, cross powerspectrum averaging means, inverse fast Fourier transform means, anddeinterleave means.

[0040] The interleave means rearranges the baseband signal digitized bythe A/D converter into N sequences sampled at chip intervals. The N fastFourier transform means pick out the N received signal sequencesrearranged by the interleave means with mutually overlapped FFT windowsof a predetermined time length and perform fast Fourier transform forthe picked out received signal sequences. The reference signal storagemeans is provided for each channel and stores a signal sequence producedby picking out a predetermined code sequence with FFT windows of a fixedtime length and fast Fourier transforming the picked out code sequenceas a reference signal. The N cross power spectrum calculation means areprovided for each channel and determine the product of the receivedsignal fast Fourier transformed by the fast Fourier transform means anda complex conjugate number of the reference signal stored in thereference signal storage means for each of the FFT windows to determinecross power spectra between the received signal and the predeterminedcode sequence. The N cross power spectrum averaging means are providedfor each channel and average the cross power spectra for the respectiveFFT windows.

[0041] The N inverse fast Fourier transform means are provided for eachchannel, and inverse fast Fourier transform the N cross power spectraaveraged by the cross power spectrum averaging means to convert thecross power spectra into N cross-correlation coefficients and output theN cross-correlation coefficients.

[0042] The deinterleave means is provided for each channel, andrearranges the N cross-correlation coefficients output from therespective inverse fast Fourier transform means in order of time toproduce and output a single cross-correlation coefficient.

[0043] The cross-correlation coefficient averaging unit averages thecross-correlation coefficients output from the cross-correlationcalculating unit over a fixed period of time. The peak detecting unitdetects one or a plurality of peaks for each channel from thecross-correlation coefficient averaged by the cross-correlationcoefficient averaging unit and outputs a timing at which the peak oreach of the peaks is obtained as a path timing.

[0044] The present aspect of the invention corresponds to a path searchcircuit which is applied to a base station apparatus which must detectpath timings of a plurality of channels (received signals from aplurality of mobile stations) simultaneously. According to the presentaspect of the invention, it is required to perform fast Fouriertransform operation for a received signal which requires the greatestamount of arithmetic operation only once irrespective of the number ofreception channels.

[0045] Therefore, the amount of arithmetic operation required perchannel can be reduced significantly.

[0046] The above and other objects, features and advantages of thepresent invention will become apparent from the following descriptionwith reference to the accompanying drawings which illustrate examples ofthe present invention.

BRIEF DESCRIPTION OF THE DRAWINGS

[0047]FIG. 1 is a block diagram showing a configuration of atransmitter-receiver which uses a path search circuit of a firstembodiment of the present invention;

[0048]FIG. 2 is a block diagram showing a configuration ofcross-correlation coefficient calculating unit 104 shown in FIG. 1;

[0049]FIG. 3 is timing chart illustrating operation of interleave unit201 shown in FIG. 2;

[0050]FIG. 4 is a timing chart illustrating a method of producing areference signal to be stored in advance in reference signal storageunit 203 shown in FIG. 2;

[0051]FIG. 5 is a timing chart illustrating operation of fast Fouriertransformers 202 ₁, 202 ₂, cross power spectrum calculating units 204 ₁,204 ₂ cross power spectrum averaging units 205 ₁, 205 ₂, and inversefast Fourier transformers 206 ₁, 206 ₂ shown in FIG. 2;

[0052]FIG. 6 is a timing chart illustrating operation of deinterleaveunit 207 and interpolation unit 208 shown in FIG. 2;

[0053]FIG. 7 is a block diagram showing a configuration ofcross-correlation coefficient calculating unit 704 of a path searchcircuit of a second embodiment of the present invention;

[0054]FIG. 8 is a block diagram showing a configuration of atransmitter-receiver which uses a path search circuit of a thirdembodiment of the present invention; and

[0055]FIG. 9 is a block diagram showing a configuration ofcross-correlation coefficient calculating unit 401 shown in FIG. 9.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS First Embodiment

[0056] A transmitter-receiver which uses a path search circuit of afirst embodiment of the present invention is a transmitter-receiverwhich uses a direct sequence code division multiple access (DS-CDMA)communication method, and includes antenna 101, and a receiver and atransmitter. The receiver includes radio receiving unit 102, A/Dconverter 103, cross-correlation coefficient calculating unit 104,cross-correlation coefficient averaging unit 105, peak detecting unit106, despreading unit 107 and demodulator 108. The transmitter includesradio transmitting unit 109, D/A converter 110, spreading unit 111, andmodulator 112.

[0057] The receiver includes a path search circuit that is composed ofradio receiving unit 102, A/D converter 103, cross-correlationcoefficient calculating unit 104, cross-correlation coefficientaveraging unit 105 and peak detecting unit 106.

[0058] Radio receiving unit 102 performs filtering and frequencyconversion matching with a chip waveform of a transmission signal for aradio signal received by antenna 101 to convert the radio signal into abaseband signal. A/D converter 103 samples the baseband signal producedby radio receiving unit 102 at a sampling rate equal to N times the chiprate to convert the baseband signal into digital received signal rxd.Cross-correlation coefficient calculating unit 104 calculatescross-correlation coefficient prof between received signal rxd digitizedby A/D converter 103 and receiving spread code rxcd that is apredetermined code sequence. Cross-correlation coefficient averagingunit 105 averages cross-correlation coefficient prof determined bycross-correlation coefficient calculating unit 104 for a fixed period oftime. Peak detecting unit 106 detects one peak or a plurality of peaksfrom the cross-correlation coefficient averaged by cross-correlationcoefficient averaging unit 105 and outputs a timing at which each peakis obtained as path timing tmg.

[0059] Despreading unit 107 despreads received signal rxd with receivingspread code rxcd allocated to the pertaining channel at the one orplural path timings determined as above. Demodulator 108 synchronouslydetects the despread received signal using a pilot signal as a referencesignal, for example, to detect reception information rxinf.

[0060] As hereinafter described, the sampling frequency of A/D converter103 may be determined in accordance with a radio bandwidth oftransmission-received signals irrespective of the required accuracy forpath timing tmg. In particular, the sampling frequency of A/D converter103 is set to a value that is higher than a Nyquist frequency (=radiobandwidth) so that waveform information may not be lost and that isequal to an integral number of times the chip rate so that laterprocessing may be simplified. Since usually a transmission-receptionfilter of a roll-off rate lower than 100% is used (radio bandwidth/chiprate≦2), a sampling frequency equal to twice the chip rate should beused for sampling. Therefore, the present embodiment will be describedin connection with a case for N=2.

[0061] First, operation of the transmitter-receiver shown in FIG. 1 willbe described. First, operation of the receiver will be described.

[0062] Referring to FIG. 1, in the present transmitter-receiver, asignal received by antenna 101 is filtered with a bandwidth matchingwith a transmission signal and frequency converted into a basebandsignal by radio receiving unit 102, and then converted into a digitalbaseband signal (rxd, represented by a complex number whose real partindicates an in-phase component and whose imaginary part indicates anorthogonal component) by A/D converter 103. Receive baseband signal rxdis supplied to cross-correlation coefficient calculating unit 104 inwhich it is used for path search and to despreading unit 107 in which itis used for despreading and demodulation.

[0063] The path search is started by cross-correlation coefficientcalculating unit 104 which first calculates cross-correlationcoefficient prof between received signal rxd and the pilot signal spreadwith receiving spread code rxcd of the pertaining channel within a rangeof a predetermined delay time.

[0064] In a cellular system, the radius of a cell covered by one basestation is prescribed, and therefore, path search should be performedwithin the range of a propagation delay of radio waves between a basestation and a mobile station when the mobile station moves within thecell. Accordingly, the delay range for determination of across-correlation coefficient can be determined in accordance with thecell radius in advance.

[0065] Cross-correlation coefficient averaging unit 105 performsin-phase averaging of cross-correlation coefficient prof within a timerange within which the phase variation of the propagation path is small,but, after a time interval within which the phase variation cannot beignored, determines power of each correlation coefficient (complexvalue) and performs power averaging. Since the cross-correlationcoefficient averaged in this manner represents a delay profile of themulti-path propagation paths, path timing tmg of the multi-pathpropagation paths can be detected by peak detecting unit 106 whichdetermines one or a plurality of peaks of the cross-correlationcoefficient (power value).

[0066] The in-phase averaging can be performed in the time domain bycross-correlation coefficient averaging unit 105 or may be performed inthe frequency domain by cross power spectrum averaging unit 205 which ishereinafter described. Where the averaging is performed in the frequencydomain, the total amount of arithmetic operation can be reduced becausethe number of times of inverse fast Fourier transform operation andinterpolation processing is comparatively small.

[0067] Despreading unit 107 despreads received signal rxd with receivingspread code rxcd allocated to the pertaining channel at the one orplural path timings determined as above. Demodulator 108 synchronouslydetects the despread received signal using the pilot signal as areference signal, for example, to detect reception information rxinf. Inthe case of the multi-path propagation paths, rake reception in whichsignals are despread and synchronously detected for each of themulti-paths are synthesized at a maximum ratio is also performed bydemodulator 108.

[0068] When transmission power control essentially required by theDS-CDMA is performed, demodulator 108 must estimate asignal-to-interference power ratio (SIR) of the received signal, comparethe estimated SIR with a target SIR necessary to achieve a predeterminedreception quality, and calculate an uplink transmission power controlcommand (Tpe_UL) for adjusting the transmission power of the other party(mobile station). Demodulator 108 further has a function of demodulatinga down transmission power control command (Tpc_DL) sent thereto from theother party (mobile station) in order to control the transmission powerof the transmitter of the transmitter-receiver.

[0069] Although such transmission power control as described above is atechnique essentially required for the DS-CDMA, no further descriptionof it is given because it is the prior art that does not directly relateto the path search circuit of the present embodiment.

[0070] Next, operation of the transmitter will be described.Transmission information txinf and uplink transmission power controlcommand Tpc_UL are multiplexed and QPSK modulated, for example, bymodulator 112 and then spread with transmitting spread code txcd byspreading unit 111. Spread transmission signal txd is converted into ananalog signal by D/A converter 110, band limited by a roll-off filter inradio transmitting unit 109, and it is frequency converted into a radiofrequency signal and then transmitted from antenna 101. The transmissionpower when the analog signal is amplified by radio transmitting unit 109is adjusted in accordance with an instruction of down transmission powercontrol command Tpc_DL.

[0071] Also such a transmitter as described above is the prior art thatdoes not directly relate to the path search circuit of the presentembodiment, and therefore, no further description thereof is given.

[0072] The path search circuit of the present embodiment features theconfiguration of cross-correlation coefficient calculating unit 104which calculates cross-correlation coefficient prof from received signalrxd and receiving spread code rxcd, and the components thereof otherthan cross-correlation coefficient calculating unit 104 can be realizedbased on the prior art.

[0073] Next, the configuration of cross-correlation coefficientcalculating unit 104 which is the characteristic part of the path searchcircuit of the present embodiment will be described with reference toFIG. 2.

[0074] As shown in FIG. 2, cross-correlation coefficient calculatingunit 104 is comprised of interleave unit 201, fast Fourier transform(FFT) units 202 ₁, 202 ₂, reference signal storage unit 203, cross powerspectrum calculating units 204 ₁, 204 ₂, cross power spectrum averagingunits 205 ₁, 205 ₂, inverse fast Fourier transform (IFTT) units 206 ₁,206 ₂, deinterleave unit 207, and interpolation unit 208.

[0075] Interleave unit 201 performs a process of rearranging digitizedreceived signal rxd into two series of received signals rxd1, rxd2sampled at time intervals of one chip which is a unit of the spreadcode.

[0076] Fast Fourier transformers 202 ₁, 202 ₂ pick out two receivedsignals rxd1, rxd2, which have been re-arranged into sequences of onechip interval by interleave unit 201, with FFT windows of apredetermined time length which overlap with each other and perform fastFourier transform for picked out received signals rxd1, rxd2.

[0077] Reference signal storage unit 203 has a signal sequence stored inadvance therein which has been produced by picking out a predeterminedcode sequence with a FFT window of a predetermined time length and fastFourier transforming the picked out code sequence. Cross power spectrumcalculating units 204 ₁, 204 ₂ calculate the products of the tworeceived signals fast Fourier transformed by fast Fourier transformers202 ₁, 202 ₂ and a complex conjugate of the reference signal stored inreference signal storage unit 203 to thereby calculate cross powerspectra between received signal rxd and receiving spread code rxcd whichis the predetermined code sequence.

[0078] Cross power spectrum averaging units 205 ₁, 205 ₂ average thecross power spectra calculated by cross power spectrum calculating units204 ₁, 204 ₂ for each FFT window, respectively.

[0079] Inverse fast Fourier transformers 206 ₁, 206 ₂ fast Fouriertransform the two values of the cross power spectra averaged by crosspower spectrum averaging units 205 ₁, 205 ₂ to convert them into twocross-correlation coefficients of a one-chip interval and output thecross-correlation coefficients.

[0080] Deinterleave unit 207 rearranges the two cross-correlationcoefficients output from inverse fast Fourier transformers 206 ₁, 206 ₂in order of time to produce one cross-correlation coefficient.Interpolation unit 208 interpolates the cross-correlation coefficientsrearranged in order of time by deinterleave unit 207 to producecross-correlation coefficient prof sampled with a required accuracy andoutputs cross-correlation coefficient prof.

[0081] Next, operation of cross-correlation coefficient calculating unit104 shown in FIG. 2 will be described in detail with reference to thetiming charts of FIGS. 3, 4, 5 and 6.

[0082]FIG. 3 is a timing chart illustrating operation of interleave unit201 which divides received signal rxd into a plurality of sequences of aone-chip interval. Received signal rxd is represented by a complexnumber whose real part indicates an in-phase component and whoseimaginary part indicates an orthogonal component. However, for the sakeof simplicity, received signal rxd is represented only by the real part.When received signal rxd is sampled with a frequency equal to N timesthe chip rate, interleave unit 201 divides received signal rxd into Nsequences of received signals sampled at one-chip intervals. Since thebandwidth/chip rate is usually lower than 2, the present embodiment willbe described in connection with a case wherein received signal rxd issampled with a frequency equal to twice the chip rate. In FIG. 3,received signal rxd is divided into sequence rxd1 composed ofeven-numbered sample values and sequence rxd2 composed of odd-numberedsample values extracted from received signal rxd.

[0083]FIG. 4 illustrates a calculation procedure of reference signal Refto be stored into reference signal storage unit 203. The referencesignal may be obtained with fast Fourier transform of a signal obtainedby spreading the pilot signal with receiving spread code rxcd, forexample. The signal obtained by spreading the pilot signal withreceiving spread code rxcd is a signal sequence for each one chip. Whenthe QPSK is used for the modulation and spreading, the signal sequenceis a complex signal of four phases with a fixed amplitude (±1±j).

[0084] As shown in FIG. 4, in order to produce reference signal Ref, thepilot signal is spread with receiving spread code rxcd and is dividedinto overlapped FFT windows (chip number=Nfft) similarly as in the caseof received signal rxd. However, unlike the case in which receivedsignal rxd is divided, when reference signal Ref is to be divided, “0”of a Nov chip is inserted into the overlapped portion. Since a frequencycomponent of a periodical signal which is repeated in the Nfft chipperiod is calculated in the fast Fourier transform, a cross-correlationcoefficient calculated using a cross power spectrum is calculated as across-correlation of cyclically shifted waveforms. Accordingly, anoverlapped section in which “0”s are inserted is provided so that thereference signal may not come out of the FFT window 1even when thereference signal is cyclically shifted within the range of the Novchips.

[0085] A signal obtained by spreading a predetermined signal like thepilot signal with receiving spread code rxcd can be used as referencesignal Ref. Therefore, this reference signal Ref is determined inadvance and is a signal repeated in a fixed period (1 frame=10 msperiod, for example), and consequently, it can be fast Fouriertransformed and stored into reference signal storage unit 203 before thereception is started.

[0086]FIG. 5 is a timing chart illustrating operation when receivedsignal rxd1 is divided into overlapped FFT windows and fast Fouriertransformed and then multiplied by a complex conjugate value of thereference signal to determine a cross power spectrum and then the crosspower spectrum is averaged and fast Fourier transformed to producecross-correlation coefficient prof of a one-chip interval, whereaftercross-correlation coefficient prof is outputted.

[0087]FIG. 5 illustrates a sequence rxd1 for N=2, obtained by extractingonly even-numbered samples at one-chip intervals by interleave unit 201.Sequence rxd2 obtained by extraction of only odd-numbered samples isalso subject to the quite same processing, and the description thereofis thus omitted.

[0088] Even-numbered sample value sequence rxd1 is picked out withoverlapped FFT windows (chip number=Nfft). The number (Nov) ofoverlapped chips is determined from a path search delay range whichdepends upon the cell radius. If the cell radius <2.5 km and the chiprate=3.84 Mcps, for example, since the propagation delay (both ways)between the base station-mobile station=2.5 km×⅔108 m/s (velocity ofradio waves)=16.67 μs, path search may be performed within the range of16.67 μs×3.84 Mcps=64 chips. In this instance, necessary and sufficientnumber Nov of overlapped chips is 64. It is to be noted, however, that,in FIGS. 4 and 5, overlapped chip number Nov is shown as being 4 forease of understanding.

[0089] The received signals divided with the FFT windows are subjectedto fast-Fourier transform processing and then multiplied by complexconjugate numbers of reference signals Ref1, Ref2, . . . to determinecross power spectra. Since a cross power spectrum (XPS1, XPS2, . . . )is output for each FFT window, each of the components of the Nfftsamples is averaged over all of the FFT windows (synchronous addition inthe frequency domain) as shown in FIG. 5.

[0090] The cross power spectra (XPS_AVG) thus averaged are subjected toinverse-fast-Fourier transform (IFFT) to determine cross-correlationcoefficients prof1 with a one-chip interval. Cross-correlationcoefficients are output over the delay range of Nfft chips as a resultof the inverse fast Fourier transform. However, those cross-correlationcoefficients outside the range of Nov chips necessary for path searchare unnecessary and therefore need not be calculated.

[0091]FIG. 6 illustrates operation of cross-correlation coefficientcalculating unit 104 in an example of double oversampling (N=2) whencross-correlation coefficients prof1 calculated from even-numberedsample value sequence rxd1 and cross-correlation coefficients prof2calculated from odd-numbered sample value sequence rxd2 are rearrangedin order of the delay time by deinterleave unit 207 and then thesampling accuracy is further raised by twice (M=2) by interpolation unit208 to detect a path timing with an accuracy in time of the totaling ¼chip (=1/(N×M)).

[0092] Referring to FIG. 6, cross-correlation coefficients (prof1)determined from an even-numbered sample value sequence (rxd1) andcross-correlation functions (prof2) determined from an odd-numberedsample value sequence (rxd2) are rearranged alternately into a singlecross-correlation sequence (prof) of a 1/N chip interval (N=2) like anoriginal received signal sequence by the deinterleave means (207). Asdescribed hereinabove, although cross-correlation coefficients of Nfftchips are outputted as a result of inverse fast Fourier transform, thosefor other than Nov chips necessary for path search may be abandoned.

[0093] Where interpolation is used to raise the accuracy of a pathtiming, “0” should be inserted between sample values before theinterpolation to raise the sampling frequency and then high frequencycomponents should be cut with a low-pass filter (LPF).

[0094] The cross-correlation coefficients (complex numbers) determinedin this manner are converted into power values by calculation of the sumof the square of the real part and the square of the imaginary part andthen averaged with respect to time to determine a delay profile of thepropagation path by cross-correlation coefficient averaging unit 105.Peak detecting unit 106 detects one or a plurality of peaks of the delayprofile to detect path timing tmg of the multi-path propagation paths.

[0095] As described above, a received signal is divided into a pluralityof N sequences at one-chip intervals and then necessary processing suchas fast Fourier transform is performed for the sequences. Now, it willbe described that the method described can decrease the amount ofcalculation.

[0096] As described hereinabove with reference to FIGS. 5 and 4, inorder to use fast Fourier transform to calculate cross-correlationcoefficients, it is necessary to use FFT windows which overlap with eachother over a time which corresponds to a delay range of thecross-correlation coefficients to be determined. This is because, sincethe fast Fourier transform assumes a periodical waveform as describedhereinabove, it is intended to prevent a cyclically shifted waveformfrom coming out of a FFT window.

[0097] Where a double oversampled receive waveform is fast Fouriertransformed as it is, in order to assure an overlap for Nov chips, anoverlap by Nov×2 samples must be taken, and therefore, the overlap rateis Nov×2/Nfft. On the other hand, if a receive waveform is divided intoa plurality of sequences at one-chip intervals as in the presentinvention, the overlap rate is Nov/Nfft, and the efficiency drop can bereduced to 1/N.

[0098] Although the overlap rate can be improved by using greater FFTwindows, since the amount of arithmetic operation in fast Fouriertransform increases in proportion to Nfft×log(Nfft), an increase in sizeof the FFT windows increases the amount of arithmetic operation in fastFourier transform at a rate corresponding to log(Nfft). Further, thisgives rise to a problem that the dynamic range of a result of arithmeticoperation halfway becomes large and the accuracy in arithmetic operation(bit width required for calculation) must be increased.

[0099] Meanwhile, since a signal obtained by fast Fourier transform of acode of a one-chip interval can be used commonly as a reference signalfor all sequences of the received signal, the storage capacity necessaryto store the reference signal can be reduced approximately to 1/N bydivision of the received signal into N series.

[0100] As a particular example, the storage capacities where the FFTwindows=256 and the delay range Nov for determination of a delayprofile=64 chips are set for a pilot signal of 1,536 chips (8symbols×256 chips) are compared.

[0101] Where a double oversampled received signal is fast Fouriertransformed as it is as in the prior art, since the FFT windows need beoverlapped over 64 chips (128 samples) with each other to perform fastFourier transform, the overlap rate=128/256=50%. In order to performfast Fourier transform for 1,536 chips, the pilot signal must be dividedinto 1,536×2/(256−128)=24 FFT windows and arithmetic operation such asfast Fourier transform must be performed with the 24 FFT windows.

[0102] On the other hand, where a received signal is divided into twoseries of even-numbered samples and odd-numbered samples to perform fastFourier transform as in the present embodiment, if FFT windows areoverlapped over 64 chips, then the overlap rate=64/256=25%. Then, thepilot signal is divided into 1,536/(256−64)=8 FFT windows for each ofthe two series, and therefore, totaling 16 times of arithmetic operationsuch as fast Fourier transform should be performed.

[0103] In this manner, between the method according to the presentembodiment wherein a digitized received signal is divided into sequencesat one-chip intervals and processing is performed for each of thesequences and the conventional method wherein a digitized receivedsignal is processed as it is, the ratio in amount of arithmeticoperation is approximately 24:16. In other words, it can be recognizedthat use of the method according to the present embodiment can achievean effect of the reduction of the amount of arithmetic operation byapproximately 33%.

[0104] In this manner, according to the present embodiment, an advantagecan be anticipated that arithmetic operation processing ofcross-correlation coefficients essentially required for path search canbe reduced by dividing a received signal into a plurality of FFT windowsto perform fast Fourier transform and performing multiplication by areference signal and averaging in the frequency domain.

[0105] Particularly since the number of chips per FFT window can beincreased by dividing a received signal into a plurality of sequences atone-chip intervals to perform fast Fourier transform, the overlapbetween FFT windows can be reduced. Since this decreases the number ofFFT windows per received signal for a fixed time length, an advantagecan be anticipated that the amount of arithmetic operation can bereduced.

[0106] If it is desired to raise the accuracy of a path timing higherthan that of a sampling period of A/D converter 103, A/D conversion isperformed at the possible lowest sampling rate and cross-correlationcoefficients or cross power spectra are determined, and theninterpolation is performed in the time domain with a required timeaccuracy. This makes it possible to suppress the amount of arithmeticoperation in fast Fourier transform operation, which involves a verygreat amount of arithmetic operation, to the possible lowest amount.

[0107] In the W-CDMA, the roll-off factor of the transmission/receptionfilters is 0.22, and therefore, A/D converter 103 should performsampling with a sampling rate equal to twice (N=2) the chip rate.Therefore, in the present embodiment, N by which received signal rxd isdivided by interleave unit 201 is 2. Where the path timing detectionaccuracy is ¼ chip, the characteristic deterioration can be suppressedto 0.25 dB or less. Therefore, cross-correlation coefficients determinedby double oversampling, for example, should be increased to twice byinterpolation (M=2) to determine a delay file with a ¼ chip accuracy,and then peaks of the delay profile should be detected to detect pathtimings.

[0108] In the double oversampling (N=2), two sequences of aneven-numbered sample value sequence and an odd-numbered sample valuesequence are output from interleave unit 201. Accordingly, fast Fouriertransformers 202 ₁, 202 ₂, cross power spectrum calculating units 204 ₁,204 ₂, cross power spectrum averaging units 205 ₁, 205 ₂ and inversefast Fourier transformers 206 ₁, 206 ₂ must be prepared individually forthe two sequences. However, since merely the same processing is repeatedtwice, the same hardware should be used to perform time divisionprocessing. Particularly such processing as described above is performedadvantageously by a digital signal processor (DSP), andcross-correlation coefficient calculating unit 104, cross-correlationcoefficient averaging unit 105 and peak detecting unit 106 can beimplemented as firmware of a DSP.

Second Embodiment

[0109] Next, a path search circuit of a second embodiment of the presentinvention will be described with reference to the drawings.

[0110] In order to raise the accuracy of a path timing higher than thesampling period of A/D converter 103, the first embodiment describedabove uses a method wherein cross-correlation coefficients areinterpolated in the time domain to raise the accuracy of path timings.

[0111] To this end, cross-correlation coefficient calculating unit 104shown in FIG. 2 includes interpolation unit 208 which oversamples anoutput of deinterleave unit 207 with a required accuracy and passes theoversampled output through a low-pass filter (LPF).

[0112] The present embodiment uses another method wherein, in order toraise the accuracy of path timings higher than the sampling period ofA/D converter 103, cross-correlation coefficients are inverse fastFourier transformed after the bandwidth of a cross power spectrum isincreased in the frequency domain.

[0113] A configuration of cross-correlation coefficient calculating unit704 used in the path search circuit of the present embodiment is shownin FIG. 7. The path search circuit of the present embodiment replacescross-correlation coefficient calculating unit 104 in the path searchcircuit of the first embodiment shown in FIG. 1 with cross-correlationcoefficient calculating unit 704. When compared with cross-correlationcoefficient calculating unit 104 shown in FIG. 2, cross-correlationcoefficient calculating unit 704 eliminates inverse fast Fouriertransformers 206 ₁, 206 ₂, deinterleave unit 207, and interpolation unit208, but additionally includes cross power spectrum converters 301 ₁,301 ₂, 303, adder 302, and inverse fast Fourier transformer 304 for aconverted cross power spectrum. In FIG. 7, like components to those inFIG. 2 are denoted by like reference characters, and overlappingdescription of them is omitted herein.

[0114] Cross power spectrum converters 301 ₁, 301 ₂ apply reflectiontwice and phase rotation in the frequency domain to average values oftwo cross power spectra obtained by cross power spectrum averaging units205 ₁, 205 ₂ and having a bandwidth equal to the chip rate to convertthe cross power spectra into a single cross power spectrum having abandwidth equal to twice the chip rate. Cross power spectrum adder 302adds the cross power spectra obtained by the conversion of cross powerspectrum converters 301 ₁, 301 ₂.

[0115] Cross power spectrum converter 303 adds a number of “0”s equal toN (=2)×(M−1) times the chip rate to a high frequency band of the crosspower spectrum obtained by adder 302 to determine a cross power spectrumhaving a bandwidth equal to N×M times the chip rate. In short, crosspower spectrum converter 303 expands the bandwidth to that correspondingto a required accuracy for path timings.

[0116] Inverse fast Fourier transform (IFFT) unit 304 inverse fastFourier transforms the cross power spectrum obtained by cross powerspectrum converter 303 and having a bandwidth increased to M times todetermine cross-correlation coefficients.

[0117] Since a receive sequence oversampled to N times is the sum of Nseries of a one-chip interval, if a cross power spectrum when each of Nsequences is sampled to N times is determined and the sum of theresulting N cross power spectra is calculated, then a cross powerspectrum of the receive sequence oversampled to N times can bedetermined.

[0118] A cross power spectrum when each sequence is oversampled to Ntimes can be determined by repeating the cross power spectrum sampled toone time by N times in the frequency domain.

[0119] Since the N sequences are sample sequences displaced successivelyby a time equal to 1/N chip, processing of delaying the time by 1/N maybe performed in the frequency domain. If the chip period is representedby Tc, a process of delaying the time by n×TC/N, for n=1, . . . N−1, isequivalent to applying to the cross power spectrum a phase rotationgiven by

exp(−j×2πk×n/N×Nfft).

[0120] Where, k=0, 1, . . . , N×Nfft−1 is a suffix indicative of a kthsample value of the cross power spectrum.

[0121] Particularly for N=2, a phase rotation given by

exp(−j×πk/Nfft), k=0, 1, . . . , 2Nfft−1

[0122] may be applied to the cross power spectrum of the odd-numberedsample value sequence.

[0123] Cross power spectrum converter 303 may pack “0”s into a highfrequency region of the cross power spectrum. In particular, in order toraise the time resolution of cross-correlation coefficients to M times,N×Nfft×(M−1) “0”s should be added to N×Nfft samples of the outputs ofcross power spectrum converters 301 ₁, 301 ₂ to obtain N×Nfft×M samples.

[0124] Inverse fast Fourier transform may be performed for the crosspower spectrum of N×Nfft×M samples, and at this time, cross-correlationcoefficient prof oversampled at 1/N×M chip intervals can be obtained.

[0125] Similar advantages to those achieved by the path search circuitof the first embodiment can be achieved by the path search circuit ofthe present embodiment, and therefore, overlapping description isomitted.

Third Embodiment

[0126] Next, a path search circuit of a third embodiment of the presentinvention will be described.

[0127] The path search circuit of the present embodiment corresponds tothe path search circuit of the first embodiment shown in FIG. 1 which isapplied to a base station apparatus which must detect path timings of aplurality of channels (received signals from a plurality of mobilestations) simultaneously.

[0128] A transmitter-receiver which uses the path search circuit of thepresent embodiment is shown in FIG. 8, and cross-correlation coefficientcalculating unit 401 shown in FIG. 8 is shown in FIG. 9.

[0129] As shown in FIG. 8, the transmitter-receiver which uses the pathsearch circuit of the present embodiment includes antenna 101, radioreceiving unit 102, A/D converter 103, cross-correlation coefficientcalculating unit 401, peak detecting unit 106 ₁, 106 ₂, despreadingunits 107 ₁, 107 ₂, demodulators 108 ₁, 108 ₂, modulators 112 ₁, 112 ₂,spreading units 111 ₁, 111 ₂, gain controlled amplifiers 113 ₁, 113 ₂,combining unit 114, radio transmitting unit 109, and D/A converter 110.

[0130] Referring to FIG. 8, while the transmitter-receiver includestransmitters and receivers for a plurality of channels,cross-correlation coefficient calculating unit 401 for detecting a pathtiming of each channel are common to all of the channels.

[0131] As shown in FIG. 9, cross-correlation coefficient calculatingunit 401 in the present embodiment includes interleave unit 201, fastFourier transform (FFT) units 202 ₁, 202 ₂, reference signal storageunits 203 ₁, 203 ₂, cross power spectrum calculating units 204 ₁ to 204₄, cross power spectrum averaging units 205 ₁ to 205 ₄, inverse fastFourier transformers 206 ₁ to 206 ₄, deinterleave units 207 ₁, 207 ₂,and interpolation units 208 ₁, 208 ₂.

[0132] Referring to FIG. 9, cross-correlation coefficient calculatingunit 401 includes one interleave unit 201 and two fast Fouriertransformers 202 ₁, 202 ₂ independently of the number of channels. Thisis because, since the channels in the DS-CDMA are code divisionmultiplexed, a single received signal corresponds to all channels andinterleave processing by interleave unit 201 and fast Fourier transformoperation by fast Fourier transformers 202 ₁, 202 ₂ are processesperformed independently of spread codes rxcd1, rxcd2 used fordistinction of the channels.

[0133] While fast Fourier transform operation must be performedrepetitively for a plurality of FFT windows, inverse fast Fouriertransform may be performed only once after averaging. Processing ofcross power spectrum calculating units 204 ₁ to 204 ₄ and cross powerspectrum averaging units 205 ₁ to 205 ₄ is multiplication or addition tobe performed once for one sample, the amount of arithmetic operation forthe processing is smaller than that of the fast Fourier transformoperation. Accordingly, it is required to perform fast Fourier transformoperation for a received signal which requires the greatest amount ofarithmetic operation only once irrespective of the number of receptionchannels, and therefore, the amount of arithmetic operation required perchannel can be reduced significantly.

[0134] When the base station apparatus receives a plurality of channelssimultaneously, the fast Fourier transform means for which the greatestamount of arithmetic operation is required can be used commonly for allchannels, and therefore, there is an advantage that the amount ofarithmetic operation per channel can be reduced.

[0135] While preferred embodiments of the present invention have beendescribed using specific terms, such description is for illustrativepurposes only, and it is to be understood that changes and variationsmay be made without departing from the spirit or scope of the followingclaims.

What is claimed is:
 1. A path search circuit for a receiver which uses aDS-CDMA communication method, for detecting a path timing, which is atiming at which spreading is performed on the transmission side, from areceived radio signal, said circuit comprising: a radio receiving unitfor filtering and frequency converting the received radio signal toconvert the received radio signal into a baseband signal; an A/Dconverter for sampling the baseband signal at a sampling rate equal to Ntimes a chip rate to convert the baseband signal into a digital signal;a cross-correlation coefficient calculating unit including interleavemeans for rearranging the baseband signal digitized by said A/Dconverter into N sequences sampled at chip intervals, N fast Fouriertransform means for obtaining the N received signal sequences rearrangedby said interleave means with mutually overlapped FFT windows of apredetermined time length and performing fast Fourier transform for thepicked out received signal sequences, reference signal storage means forstoring a signal sequence produced by picking out a predetermined codesequence with FFT windows of a fixed time length and fast Fouriertransforming the picked out code sequence as a reference signal, N crosspower spectrum calculation means for determining the product of thereceived signal fast Fourier transformed by said fast Fourier transformmeans and a complex conjugate number of the reference signal stored insaid reference signal storage means for each of the FFT windows todetermine cross power spectra between the received signal and thepredetermined code sequence, N cross power spectrum averaging means foraveraging the cross power spectra for the respective FFT windows, Ninverse fast Fourier transform means for inverse fast Fouriertransforming the N cross power spectra averaged by said cross powerspectrum averaging means to convert the cross power spectra into Ncross-correlation coefficients and outputting the N cross-correlationcoefficients, and deinterleave means for rearranging the Ncross-correlation coefficients output from said respective inverse fastFourier transform means in order of time to produce and output a singlecross-correlation coefficient; a cross-correlation coefficient averagingunit for averaging the cross-correlation coefficients output from saidcross-correlation calculating unit over a fixed period of time; and apeak detecting unit for detecting one or a plurality of peaks from thecross-correlation coefficient averaged by said cross-correlationcoefficient averaging unit and outputting a timing at which the peak oreach of the peaks is obtained as a path timing.
 2. A path search circuitaccording to claim 1 , wherein the value N is an integer lowest amongvalues which satisfy relationship: N≧radio bandwidth/chip rate.
 3. Apath search circuit according to claim 2 , further comprisinginterpolation means for oversampling the N cross-correlation functionsoutput from said deinterleave means as sequences of a time intervalequal to 1/N the chip interval to M times, where M is a positiveinteger, and passing the oversampled cross-correlation coefficientsthrough a low-pass filter to produce cross-correlation coefficientsoversampled to N×M times the chip rate and then outputting the producedcross-correlation coefficients.
 4. A path search circuit for a receiverwhich uses a DS-CDMA communication method for detecting a path timing,which is a timing at which spreading is performed on the transmissionside, from a received radio signal, comprising: a radio receiving unitfor filtering and frequency converting the received radio signal toconvert the received radio signal into a baseband signal; an A/Dconverter for sampling the baseband signal at a sampling rate equal to Ntimes a chip rate to convert the baseband signal into a digital signal;a cross-correlation coefficient calculating unit including interleavemeans for rearranging the baseband signal digitized by said A/Dconverter into N sequences sampled at chip intervals, N fast Fouriertransform means for picking out the N received signal sequencesrearranged by said interleave means with mutually overlapped FFT windowsof a predetermined time length and performing fast Fourier transform forthe picked out received signal sequences, reference signal storage meansfor storing a signal sequence produced by picking out a predeterminedcode sequence with FFT windows of a fixed time length and fast Fouriertransforming the picked out code sequence as a reference signal, N crosspower spectrum calculation means for determining the product of thereceived signal fast Fourier transformed by said fast Fourier transformmeans and a complex conjugate number of the reference signal stored insaid reference signal storage means for each of the FFT windows todetermine cross power spectra between the received signal and thepredetermined code sequence, N cross power spectrum averaging means foraveraging the cross power spectra for the respective FFT windows, Nfirst cross power spectrum conversion means for applying reflection by Ntimes and phase rotation in the frequency domain to the N cross powerspectra averaged by said cross power spectrum averaging means and havinga bandwidth equal to the chip rate to convert the N cross power spectrainto a single cross power spectrum having a bandwidth equal to N timesthe chip rate, cross power spectrum addition means for adding the Ncross power spectra converted by each of said first cross power spectrumconversion means, second cross power spectrum conversion means foradding a number of “0”s equal to N (M−1) times the chip rate to a highfrequency of the cross power spectrum obtained by said addition means,where M is a positive integer, and inverse fast Fourier transform meansfor inverse fast Fourier transforming the cross power spectrum obtainedby said second power spectrum conversion means and having a bandwidthincreased to M times to determine a cross-correlation coefficient; across-correlation coefficient averaging unit for averaging thecross-correlation coefficients output from said cross-correlationcalculating unit over a fixed period of time; and a peak detecting unitfor detecting one or a plurality of peaks from the cross-correlationcoefficient averaged by said cross-correlation coefficient averagingunit and outputting a timing at which the peak or each of the peaks isobtained as a path timing.
 5. A path search circuit for a receiver whichuses a DS-CDMA communication method for detecting a path timing, whichis a timing at which spreading is performed on the transmission side,for each of a plurality of channels from a received radio signal,comprising: a radio receiving unit for filtering and frequencyconverting the received radio signal to convert the received radiosignal into a baseband signal; an A/D converter for sampling thebaseband signal at a sampling rate equal to N times a chip rate toconvert the baseband signal into a digital signal; a cross-correlationcoefficient calculating unit including interleave means for rearrangingthe baseband signal digitized by said A/D converter into N sequencessampled at chip intervals, N fast Fourier transform means for pickingout the N received signal sequences rearranged by said interleave meanswith mutually overlapped FFT windows of a predetermined time length andperforming fast Fourier transform for the picked out received signalsequences, reference signal storage means provided for each channel forstoring a signal sequence produced by picking out a predetermined codesequence with FFT windows of a fixed time length and fast Fouriertransforming the picked out code sequence as a reference signal, N crosspower spectrum calculation means provided for each channel fordetermining the product of the received signal fast Fourier transformedby said fast Fourier transform means and a complex conjugate number ofthe reference signal stored in said reference signal storage means foreach of the FFT windows to determine cross power spectra between thereceived signal and the predetermined code sequence, N cross powerspectrum averaging means provided for each channel for averaging thecross power spectra for the respective FFT windows, N inverse fastFourier transform means provided for each channel for inverse fastFourier transforming the N cross power spectra averaged by said crosspower spectrum averaging means to convert the cross power spectra into Ncross-correlation coefficients and outputting the N cross-correlationcoefficients, and deinterleave means provided for each channel forrearranging the N cross-correlation coefficients output from saidrespective inverse fast Fourier transform means in order of time toproduce and output a single cross-correlation coefficient; across-correlation coefficient averaging unit for averaging thecross-correlation coefficients output from said cross-correlationcalculating unit over a fixed period of time; and a peak detecting unitfor detecting one or a plurality of peaks for each channel from thecross-correlation coefficient averaged by said cross-correlationcoefficient averaging unit and outputting a timing at which the peak oreach of the peaks is obtained as a path timing.
 6. A path search circuitaccording to claim 5 , wherein the value N is an integer lowest amongvalues which satisfy relationship N≧radio bandwidth/chip rate.
 7. A pathsearch circuit according to claim 5 , further comprising interpolationmeans for oversampling the N cross-correlation functions output fromsaid deinterleave means as sequences of a time interval equal to 1/N thechip interval to M times, where M is a positive integer, and passing theoversampled cross-correlation coefficients through a low-pass filter toproduce cross-correlation coefficients oversampled to N×M times the chiprate and then outputting the produced cross-correlation coefficients.